Local oscillator apparatus and method

ABSTRACT

A direct conversion satellite tuner is fully integrated on a common substrate. The integrated tuner receives an RF signal having a plurality of channels and down-converts a selected channel directly to baseband for further processing. The integrated tuner includes on-chip local oscillator generation, tunable baseband filters, and DC Offset cancellation. The integrated tuner can be implemented in a completely differential I/Q configuration for improved electrical performance. The entire direct conversion satellite tuner can be fabricated on a single semiconductor substrate using standard CMOS processing, with minimal off-chip components. The tuner configuration described herein is not limited to processing TV signals, and can be utilized to down-convert other RF signals to an IF frequency or baseband.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation application of U.S. patentapplication Ser. No. 10/647,588, filed on Aug. 26, 2003, which is adivisional application of U.S. patent application Ser. No. 09/995,695,filed on Nov. 29, 2001, which claims priority to U.S. ProvisionalApplication No. 60/250,616, filed on Nov. 29, 2000, both of which areincorporated herein by reference in its entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention generally relates to receiver circuits, and morespecifically to a direct conversion satellite tuner that is integratedon a single semiconductor substrate.

2. Background Art

Direct satellite television broadcasts television signals directly froma satellite to a user, without any terrestrial re-transmission of thetelevision signals. At the user location, a satellite dish receives thesatellite signals, and a satellite receiver retrieves the basebandinformation for display on a standard television set.

In addition to television service, direct satellite television systemsare also being configured to offer Internet service, including broadbandor high speed Internet service.

Direct satellite television signals occupy a frequency spectrum from 950MHz to 2150 MHz, with a channel spacing of approximately 29.16 MHz and achannel bandwidth of approximately 26.4 MHz. Therefore, approximately 40channels are available in the 950-2150 MHz frequency band.

The modulation scheme that is utilized for satellite television signalsis quadrature phase shift keying (QPSK). QPSK provides a dataconstellation having 4 possible positions, where each positionrepresents two data bits. As a result, more than 40 information channelscan be transmitted over the allotted frequency bandwidth since eachconstellation position represents two data bits.

Future satellite television systems may be expanded to 8 PSK, furtherincreasing the number of information channels that can be transmittedover the allotted frequency bandwidth.

Conventional satellite receivers utilize a hybrid configuration ofmultiple chips, boards, and/or substrates. For example, the localoscillator source, one or more mixers, and the baseband filter aretypically on different substrates from each other. As a result, thesemultiple substrates must be assembled and electrically connectedtogether, which increases manufacturing time and cost. Furthermore,electrical parasitics are associated with driving high frequency signalsfrom one substrate to another, and can reduce electrical performance.Often times, individual components need to be tuned to compensate forthe parasitics associated with driving a high frequency signal from onesubstrate to another.

Therefore, a single chip solution is highly desirable for satellitetelevision tuners. The single chip solution will eliminate the need toconnect multiple substrates together during manufacturing, and thereforewill lead to a reduction in manufacturing time and cost. The single chipsolution will likely improve electrical performance of the tuner as theparasitics associated with driving a signal off-chip will be eliminated.Additionally, the single chip solution will reduce the size of thesatellite tuners, which becomes more critical for non-TV setapplications. Therefore, what is required is a satellite tunerarchitecture that can be implemented on a single semiconductorsubstrate.

Additionally, the channel bandwidth requirements at baseband vary fromone service provider to another, often based on geographic location. Forexample, a European service provider will typically have a differentbandwidth requirement than a North American service provider.Conventional tuners do not have the ability to tune baseband bandwidthon chip in realtime. Therefore, the service provider must be identifiedduring manufacturing so that the baseband filter bandwidth can setaccordingly. Therefore, it would be advantageous for a single-chip tunerto have the capability of tuning the bandwidth of the baseband output sothat the tuner could be mass produced without prior knowledge of theservice provider and the corresponding baseband bandwidth requirement.

BRIEF SUMMARY OF THE INVENTION

The present invention is directed to an integrated tuner for processingradio frequency (RF) signals that have a plurality of channels. Theintegrated tuner down-converts a selected channel directly from an RFfrequency to a baseband frequency, for subsequent processing.

The integrated tuner includes a local oscillator (LO) generationcircuit, a differential direct conversion mixer, a differential tunablelowpass filter, and a DC compensation circuit, all of which are disposedon a common substrate. The LO generation circuit generates adifferential LO signal that is sent to the differential directconversion mixer. The differential direct conversion mixer circuit mixesthe RF signal with the differential LO signal, where the frequency ofthe differential LO signal is determined to down-convert a selectedchannel in the RF signal directly to baseband. The LO correction circuitadjusts the amplitude level of the differential LO signal so as toimprove the performance of the differential direct conversion mixer. Thedifferential tunable lowpass filter filters the differential basebandsignal to remove unwanted frequencies. The DC compensation circuitdetects any DC offset in the differential baseband signal and cancelsthe DC offset using negative feedback.

The integrated tuner is fully integrated on a single semiconductorsubstrate and can use standard semiconductor processes, such as CMOS. Assuch there is no need for assembling multiple different substrates orchips. Unlike conventional tuners, both the local oscillator (LO) andthe baseband filtering functions are performed on-chip, representativeof the full integration. Furthermore, the baseband filter tuning is alsoperformed on-chip. Furthermore, in embodiments of the invention, thetuner is completely differential, thereby improving phase noiseperformance and facilitating the mitigation of unwanted common modevoltages and DC offset.

In embodiments of the invention, the integrated tuner is configured toprocess an RF signal that is a direct satellite television signal thatoccupies a frequency range from 950-2150 MHz.

In embodiments of the invention, the LO generation circuit is a PLLhaving a plurality of VCOs, where each VCO covers a different frequencyband. A VCO is selected based on the desired frequency of thedifferential LO signal and the channel that is to be selectivelydown-converted to baseband.

In embodiments of the invention, the integrated tuner is configured forin-phase (I) and quadrature (Q) operation. Therefore, the LO generationcircuit generates I and Q differential LO signals. Furthermore, thedifferential direct conversion mixer includes I and Q mixers thatgenerate corresponding I and Q baseband signals. The I and Q basebandsignals are received by corresponding first and second tunable lowpassfilters, and corresponding first and second DC compensation circuits.

In embodiments of the invention, the differential direct conversionmixer includes an RF transconductance circuit and a LO switchingcircuit. The RF transconductance circuit includes a pair of field effecttransistor to convert the received differential RF signal to adifferential RF current. The differential direct conversion mixerfurther includes a means of adding a DC current (that does not flowthrough the LO switching circuit) to the pair of transistors so as tominimize flicker noise.

In embodiments of the invention, the differential tunable lowpass filterincludes a plurality of integrators, each integrator having a resistorand a capacitor. The cutoff frequency of the differential tunablelowpass filter is tuned by adjusting either the resistor or thecapacitor in the integrators. In one embodiment, the capacitor is afixed metal oxide semiconductor capacitor (MOSCAP), and the cutofffrequency of the differential lowpass filter is tuned by adjusting avalue of the resistor.

In embodiments of the invention, the DC compensation circuit includes aDC servo circuit that detects any DC offset voltage in a second basebandamplifier and cancels the DC offset voltage at the output of a firstamplifier, which can be a variable gain amplifier (VGA). The DC servocircuit includes a first transconductance amplifier, a lowpass filter,and a second transconductance amplifier. The first transconductanceamplifier detects the output of the second baseband amplifier, andconverts the output of the second baseband amplifier to a differentialcurrent. The lowpass filter has a cutoff frequency that passes only theDC offset information in the differential current and rejects thebaseband signal information. Finally, the second transconductanceamplifier is connected 180 degrees out-of-phase with the VGA so that theDC offset is canceled at the input of the VGA.

Further features and advantages of the present invention, as well as thestructure and operation of various embodiments of the present invention,are described in detail below with reference to the accompanyingdrawings.

BRIEF DESCRIPTION OF THE DRAWINGS/FIGURES

The present invention is described with reference to the accompanyingdrawings. In the drawings, like reference numbers indicate identical orfunctionally similar elements. Additionally, the left-most digit(s) of areference number identifies the drawing in which the reference numberfirst appears.

FIG. 1 illustrates a direct satellite TV environment.

FIG. 2 illustrates down-conversion of a selected channel in the spectrumassociated with the direct satellite TV environment that is described inFIG. 1.

FIG. 3 illustrates an integrated direct conversion tuner according toembodiments of the present invention.

FIG. 4 illustrates a VCO control table that identifies which amplifiersshould be powered up to select a particular VCO.

FIG. 5A illustrates a functional description of a polyphase circuit.

FIG. 5B further illustrates the LO generation circuit according toembodiments of the present invention.

FIG. 6 illustrates a modified differential Gilbert cell mixer accordingto embodiments of the invention.

FIG. 7 further describes the modified Gilbert cell mixer in FIG. 6according to embodiments of the invention.

FIG. 8 illustrates a tunable baseband lowpass filter implementingleapfrog synthesis according to embodiments of the invention.

FIG. 9 illustrates a conventional lowpass Butterworth filter that issimulated by the tunable bandpass filter in FIG. 8, according toembodiments of the present invention.

FIG. 10 illustrates an active integrator stage of tunable bandpassfilter in FIG. 8, according to embodiments of the present invention.

FIG. 11 illustrates level shifting between two active integrator stagesin the tunable bandpass filter of FIG. 8, according to embodiments ofthe present invention.

FIG. 12 illustrates a level shifting circuit, according to embodimentsof the present invention.

FIG. 13 illustrates a tunable resistor that is used to tune the cutofffrequency of the tunable filter in FIG. 8, according to embodiments ofthe present invention.

FIG. 14 illustrates a DC compensation circuit that eliminates DC offsetaccording to embodiments of the present invention.

FIG. 15 further illustrates a lowpass filter that has as an externalcapacitor in the DC servo feedback loop of the DC compensation circuit.

FIG. 16 illustrates a flowchart 1600 that describe the operation of theDC servo loop according to embodiments of the present invention.

FIG. 17 illustrates a filter tuning and compensation circuit associatedwith the lowpass baseband filter, according to embodiments of thepresent invention.

FIG. 18 illustrates a switched capacitor that is used in the filtertuning and compensation circuit of FIG. 17, according to embodiments ofthe present invention.

FIG. 19 illustrates a low noise amplifier according to embodiments ofthe present invention.

FIG. 20 illustrates a high gain stage and a low gain stage in the lownoise amplifier according to embodiments of the present invention.

FIG. 21 illustrates a unit gain amplifier that is repeated in the highgain stage the low gain stage of the low noise amplifier according toembodiments of the present invention.

FIG. 22 illustrates a matching circuit for the low noise amplifieraccording to embodiments of the present invention.

FIG. 23 illustrates a top view of a CMOS twisted pair according toembodiments of the present invention.

FIG. 24 illustrates an isometric view of a CMOS twisted pair accordingto embodiments of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

Example Environment

Before describing the invention in detail, it is useful to describe anexample satellite environment for the invention. The invention is notlimited to the satellite environment that is described herein, and isapplicable to other satellite and non-satellite applications as will beunderstood by those skilled in the relevant arts based on thediscussions given herein.

FIG. 1 illustrates a direct satellite TV environment 100. Referring toFIG. 1, a satellite 102 transmits a wireless signal 104 that carries TVprogramming information. An antenna 106 receives the wireless signal 104and generates an electrical signal 107. For satellite TV, the signal 104is exemplarily operated at 12 GHz and therefore the signal 107 is alsoat 12 GHz. The down-converter 108 receives the signal 107 anddown-converts the signal 107 to generate a signal 110, where the signal110 occupies a bandwidth of 950-2150 MHz. A tuner assembly 112 receivesthe signal 110 and down-converts a selected channel in the signal 110 tobaseband so as to produce a baseband signal 114. A baseband processor116 processes the baseband signal 114 for display on the monitor 118. Inembodiments, the monitor 118 is a TV, computer, or other display device.Furthermore, the monitor 118 can include an MPEG decoder, which is knownto those skilled in the arts.

In a typical satellite TV environment, the antenna 106 is locatedoutside so the antenna 106 has a clear line-of-site to the satellite102. The down-converter 108 (also called a low-noise block) is locateddirectly approximate to the antenna 106 so to as the immediatelydown-convert the 12 GHz signal to the lower frequency range of 950-2150MHz, and thereby minimizes signal loss at the higher frequency 12 GHzfrequency. The tuner 112 is located some distance away from the antenna106, typically in an enclosed structure (e.g. residence) and isconnected to the down-converter 108 via a coaxial cable, or otherequivalent transmission medium.

FIG. 2 further illustrates the frequency translation performed by thetuner 112. Referring to FIG. 2, the spectrum 110 contains multiplechannels 204 a-n that occupy from 950-2150 MHz. A local oscillatorsignal 206 is tuned to the center of a selected channel 204, which couldbe any of the channels 204 in the signal 110. In FIG. 2, the selectedchannel is the channel 204 c for example purposes. The tuner 112down-converts the selected channel 204 directly to baseband, orapproximately 0 Hz. In embodiments of the invention, the tuning stepsize of the LO 206 is relatively coarse, which causes the signal 114 tobe shifted off of 0 Hz, by up to 2 MHz. This frequency shift off of 0 Hzfor the signal 114 can be detected and compensated for by the basebandprocessor 116.

Integrated Tuner

FIG. 3 illustrates the tuner assembly 112 in further detail. The tunerassembly 112 includes a balun 302, and a tuner 306. The balun 302receives the signal 110, which is single-ended because the antenna 106and corresponding cable is single-ended. The balun 302 converts thesingle-ended RF signal 110 to a differential RF signal 303. The tuner306 receives the differential RF signal 303 and down-converts a selectedchannel (e.g. channel 204 c in FIG. 2) to baseband frequencies. Theselected channel is down-converted in an IQ format, represented by I andQ baseband signals 114 a and 114 b. The operation of the tuner 306 isdescribed in further detail below.

The tuner 306 is fully integrated on a single semiconductor substrateusing standard semiconductor processes, such as CMOS. As such there isno need for assembling multiple different semiconductor substrates.Unlike conventional tuners, both the local oscillator (LO) and thebaseband filtering functions are performed on-chip, representative ofthe full integration. Furthermore, the bandpass filter tuning is alsoperformed on-chip. Furthermore, in embodiments of the invention, thetuner 306 is completely differential, thereby improving phase noiseperformance and facilitating the mitigation of unwanted common modevoltages and DC offset.

The tuner 306 includes a LNA 304, a LO generator 308, a LO correctioncircuit 313, IQ mixers 334 a and 334 b, fixed baseband filters 336 a and336 b, and DC offset compensation circuits 348 a and 348, and tunablebaseband filters 344 a and 344 b. One or more of these components can betuned using an I²C interface 354. The tuner 306 is further described asfollows.

The LNA 304 receives the differential RF signal 303 from the off-chipbalun 302. The LNA 304 variably amplifies the differential RF signal 303according to an RF AGC control 352, to produce a differential RF signal305. The differential RF signal 305 is forwarded to the RF input of theIQ mixers 334 for down-conversion.

The LO generator 308 generates an in-phase (I) local oscillator (LO)signal 206 a and a quadrature (Q) local oscillator (LO) signal 206 b,collectively referred to as the LO signals 206. As will be understood,the LO signals 206 a and 206 b have a quadrature phase relationship andare differential. The frequency of the LO signals 206 is tuned todown-convert the selected channel in the signal 110 to baseband. Asshown in FIG. 2, the frequency of the LO 206 is determined so that it issubstantially in the middle of the channel 204 that is to bedown-converted to baseband. The frequency of the LO signals 206 can beexternally controlled using the I²C interface 354 to adjust the outputfrequency of a PLL 316 in the LO generation 308, as will be discussedfurther below.

LO correction circuit 313 receives the LO signals 206 and adjusts theamplitude to a predetermined level, so as to produce LO signals 333. Theamplitude of the LO signals 333 is adjusted to improve or maximize theelectrical performance of IQ mixers 334. More specifically, amplifiers330 a and 330 b variable amplify the respective LO signals 206 a and 206b according to a feedback signal from the level detect 332. The leveldetect 332 detects the amplitude level at the output of the variableamplifiers 330, and generates the feedback signal that controls the gainof the amplifiers 330, so as to produce a desired peak-to-peak voltagelevel at the output of the amplifiers 330. The peak-to-peak voltagelevel is externally controlled via the I²C interface 354. In oneembodiment, the peak-to-peak voltage is set to 1 volt peak-to-peak, butother peak-to-peak voltages could be chosen.

In one embodiment, each amplifier 330 includes multiple field effecttransistor (FET) amplifiers. The feedback signal from the level detect332 controls the current of at least one of the FET amplifiers so as tocontrol the gain of the respective FET amplifier, and thereby controlsthe signal level of the corrected LO signals 333.

IQ mixers 334 receive the corrected LO signals 333 and differential RFsignal 305. Mixer 334 a mixes the differential RF signal 305 with the Icorrected LO signal 333 a, to produce an I baseband signal 335 a. Mixer335 b mixes the differential RF signal 305 with a Q corrected LO signal333 b, to produce a Q baseband signal 335 b. As stated above, thefrequency of the LO signals 333 is configured so that a selected channelin the differential RF signal 305 is down-converted directly tobaseband. In one embodiment, each of the mixers 334 is a modifieddifferential Gilbert cell mixer, the configuration and operation ofwhich will is described further in following sections.

In addition to generating the baseband signal, the mixers 334 alsoup-convert the RF signal 305 to a sum frequency, which falls atf_(LO)+f_(RF). The sum frequency occurs at approximately twice thefrequency f_(LO), since the f_(LO) is selected to be equal to theselected channel in the RF signal 305. The lowpass filters 336 a and 336b substantially reject the sum frequency in the respective I basebandsignal 335 a and the Q baseband signal 335 b. However, the lowpassfilters 336 a,b pass the baseband frequencies (f_(RF)−f_(LO)) from therespective I and Q signals 335 a and 335 b, to produce I and Q basebandsignals 337 a and 337 b. In embodiments, the filters 336 also removesome of the unwanted channels that were not down-converted to baseband,which produces lower distortion in stages that follow.

Baseband AGC amplifiers 338 a,b receive the respective I and Q basebandsignals 337 a,b. The AGC circuit 338 a variably amplifies the I basebandsignal 337 a according to a baseband AGC signal 350, to produce abaseband signal 343 a. Likewise, the AGC circuit 338 b variablyamplifies the Q baseband signal 337 b according to the baseband AGCsignal 350, to produce a baseband signal 343 b.

The RF AGC 304 and the baseband AGC 338 operate as a dual AGC to controlthe amplitude of the I and Q baseband signals 114. Preferably, the RFAGC 304 is configured to maintain a low noise figure, when compared tothat of the baseband AGCs 338. When receiving an RF signal 303 having arelatively high level, the gain of the baseband AGCs 338 is reducedprior to the gain of the RF AGC 304. This is done to maximize thesignal-to-noise performance of the baseband output signals 114.

Baseband filter 344 a lowpass-filters the I baseband signals 343 a toremove unwanted frequencies, producing baseband signal 345 a. Likewise,baseband filter 344 b lowpass-filters the Q baseband signal 343 b toremove unwanted frequencies, producing baseband signal 345 b. Thebaseband filters 334 are tunable, where their lowpass cutoff frequencyis determined by a tuning control circuit 340. In embodiments of theinvention, the lowpass cutoff frequencies are tunable from 2 MHz to 36MHz. It is advantageous to have a tunable baseband filter on-chipbecause the baseband channel bandwidth for satellite televisionreception can even vary from one service provider to another. Therefore,by having a tunable baseband filter on-chip, a single tuner design canbe mass-produced without prior knowledge of the service provider(s) andtheir respective baseband channel bandwidth requirements.

Buffer amplifier 346 a receives the I baseband signal 337 a andamplifies the baseband signal 345 a to produce the I baseband signal 114a. Likewise, the amplifier 346 b receives the Q baseband signal 337 band amplifies the Q baseband signal 345 b to produce the Q basebandsignal 114 b.

As discussed herein and illustrated in FIG. 3, the tuner 306 isconfigured to be completely differential, which improves electricalperformance, including phase noise performance. However, in differentialconfigurations, DC offsets can occur between the positive and negativecomponents of the differential signal. DC offsets are caused bycomponent mismatches that are along a receiver chain. Withoutcompensation, these DC offsets can saturate the baseband components,thereby preventing proper reception of the desired channel. One way toremove DC offset is to AC couple the receiver components. For example,the RF AGCs 338 a and 338 b could be AC coupled to the respective outputof the filters 336 a and 336 b using capacitors. However, this isproblematic at baseband frequencies because the capacitors would have tobe very large and therefore hard to implement on-chip.

Hence, DC compensation circuits 348 a and 348 b remove the DC offsetsfrom the respective I and Q baseband signals 114 a and 114 b using asubtractive feedback technique, without the need for series capacitors.More specifically, a DC servo circuit 342 a senses the output of theamplifier 346 a and determines an average 339 a for any DC offset in theI baseband signal 114 a. The average DC offset 339 a is inverted and isfed back to the output of the amplifier 338 a so as to cancel the DCoffset at the output of the variable gain amplifier 338 a, therebyremoving any DC offset from I baseband signal 114 a. A DC servo circuit342 b senses the output of the amplifier 346 b and determines an average339 b for any DC offset in the Q baseband signal 114 b. The average DCoffset 339 b is inverted and is fed back to the output of the amplifier338 b so as to cancel the DC offset at the output of the variable gainamplifier 338 b, thereby removing any DC offset from Q baseband signal114 b.

Further details of the LO generation circuit 313, the IQ mixers 334, theDC compensation circuits 348, and the lowpass filters 344 are describedbelow.

Local Oscillator Generation

The LO circuit 308 includes a phase lock loop (PLL) 316 and twopolyphase circuits 310 and 312 to generate the I and Q LO signals 206 aand 206 b. The PLL 316 includes multiple VCOs 318 a-d, where one VCO 318is selected based on the desired frequency for the I and Q LO signals206 a,b. In one embodiment: VCO 318 a covers the frequency range from950-1250 MHz; VCO 318 b covers the frequency range from 1250-1550 MHz;VCO 318 c covers the frequency range from 1550-1850 MHz; and VCO 318 dcovers the frequency range from 1850-2150 MHz. The VCOs 318 are notlimited to the mentioned frequency ranges as other frequency divisionsmay be possible.

The PLL 316 operates according to PLL feedback principles. Morespecifically, a VCO output 313 c of the selected VCO 318 is fed back toa divider 314, which divides the frequency of the selected VCO 318 by1/N, to produce a divider output 315. A phase detector 324 compares thedivider output 315 of the divider 314 with a reference signal 325, andgenerates an error signal 323 that represents the phase or frequencydifference between the two signals 315 and 325. The error signal 323drives a charge pump 322 that generates an output current 321 thatdrives a loop filter 320. The loop filter 320 generate a voltageaccording to the charge pump current 321, which tunes the frequency ofthe selected VCO 318 to remove any frequency and/or phase differencebetween the output 313 and the reference signal 325. As such, the PLL316 operates as a self correcting feedback loop that corrects anyfrequency and/or phase difference between the VCO output 313 c and thereference signal 325.

The reference signal 325 is generated by a crystal oscillator 328 and adivider 326. The reference signal 325 determines when the phase detector324 updates the error signal 323, and therefore determines the frequencyresolution (or Δfreq change) in the output of the VCO output signal 313.There is an inverse relationship between frequency resolution and phasenoise in a phase lock loop. The finer the frequency resolution, thehigher the phase noise of the PLL 316. However, it is often desirable tohave fine frequency resolution in a phase lock loop to produce an outputsignal frequency that is as accurate as possible. Hence, theprogrammable divider 326 allows for the optimization between thesecompeting goals of phase noise and frequency control. In one embodiment,the oscillator 328 is a 16 Mhz oscillator, and the programmablefrequency divider 326 is capable of dividing the output of the crystaloscillator 328 by factors of either 8, 4, or 2. If the divider 326 isset to 2, then the frequency of the VCO output 313 varies in 8 MHzsteps. If the divider 326 is set to 8, then the frequency of the VCOoutput 313 varies in 2 MHz steps, allowing for finer frequency controlthan for the divide-by-two selection. However, the phase noise is worsefor the divide by 8 selection because there is a longer time periodbetween the time that the error signal 323 is updated.

As stated herein, the frequency of the local oscillator 206 is tuned toselect the channel in the RF signal 305 that is down-converted tobaseband. Once the VCO 318 is selected, the frequency of the selectedVCO 318 can be further tuned by adjusting either the reference signal325 or the divider ratio of the divider 314. The tuning instructions canbe received using the I²C bus 354 to adjust either the reference signal325, the divider 314, or some other way of tuning the PLL 308.

In addition to being fed to the divider 314, the output of the selectedVCO 318 is fed to one of the polyphase circuits 310 and 312, over lines313 a or 313 b. The polyphase circuit 310 covers the range from 950 to1550 MHz, and is connected to the VCOs 318 a and 318 b by thedifferential lines 313 b. The polyphase circuit 312 covers the rangefrom 1550 MHz to 2150 MHz, and is connected to the VCOs 318 c and 3189 dover the lines 313 a. As illustrated, multiple amplifiers 356 a-h arealso connected in series between the outputs of the VCOs 318 and thepolyphase circuits 310 and 312. For example, amplifiers 356 a-d areconnected directly at the output of the PLL 318 a-d, respectively.Furthermore, the amplifier 356 h is connected in-series between the VCOs318 a,b and the polyphase circuit 310. Likewise, the amplifier 356 g isconnected in-series between the VCOs 318 c,d and the polyphase circuit312. Furthermore, the amplifier 356 e is connected between the output ofVCOs 318 a and 318 b and the divider 314. The amplifier 356 f isconnected between the output of the VCOs 318 c and 318 d and the divider414. Furthermore, amplifiers 356 i and 356 j are connected to the outputof the respective polyphase circuits 310, 312. The selection among theVCOs 318 is discussed further below.

In one embodiment, the selected VCO 318 is determined by controlling thepower to amplifiers 356 using the I²C interface 354. In other words,some of the amplifiers 356 are turned-on and some of the amplifiers 356are turned-off to connect the desired VCO 318 to the appropriatepolyphase circuit, and to disconnect the remaining VCOs 318 from thepolyphase circuits. For example, assuming that the VCO 318 a is theselected VCO 318, then the amplifiers 356 a and 356 h are powered-up toconnect the VCO 318 a to the polyphase circuits 310. Additionally, theamplifier 356 e is also turned-on to connect the VCO 318 a to thedivider 314. Additionally, the amplifier 356 i is also powered up at theoutput of the polyphase circuit 310. The power to the remainingamplifiers 356 is turned-off and therefore no other VCO 318 (other than318 a) is connected to either the polyphase circuits 310 and 312, or tothe divider 314. FIG. 4 continues the example for each of the VCOs 318by providing a table 400 that defines the power state for the amplifiers356 that is needed to enable a specified VCOs 318.

The polyphase circuit 310 receives the output 313 b from the VCO 318 aor the VCO 318 b when one of these VCOs are active, and generates I andQ LO outputs 206 a or 206 b. Likewise, the polyphase filter 312 receivesthe output 313 a from the VCO 318 c or the VCO 318 d when one of theseare active, and the generates I and Q LO outputs 206 a or 206 b. FIG. 5further illustrates the polyphase circuits 310,312. Referring to FIG. 5a, each polyphase circuit 310,312 includes a signal divider 502 andphase shifts(s) 504 that are configured so that one output 506 of thepolyphase circuit is phase-shifted by 90 degrees with respect to theother output 508. The polyphase circuits 310, 312 operate over a limitedbandwidth as the phase-shift 504 only provides an accurate 90 degreephase-shift over a limited frequency bandwidth. Hence the phase-shift504 in the polyphase circuit 310 is optimized to operate over thefrequency range of 950 to 1550 MHz, which corresponds with the frequencycovered by the VCOs 318 a and 318 b. Similarly, the phase-shift 504 inthe polyphase circuit 312 is optimized for a frequency range of 1550 to2150 MHz, which corresponds with the frequency covered by the VCOs 318 cand 318 d. In one embodiment, the phase-shift 504 in the polyphasecircuits 310,312 is optimized so that the quadrature phase error betweenthe I and Q LO signals 206 a and 206 b is less than 2 degrees. In otherwords, the phase difference between the I and Q LO signals is 90 degrees+/−2 degrees across the designated frequency band.

FIG. 5B further illustrate path connections between the PLL 316 and thepolyphase circuits 310 and 312. Of note in FIG. 5B, is that there areswitches connected to the outputs of the VCOs 318 that are closed forthe VCO that is selected for operation. Furthermore, the polyphasecircuits 310,312 have 4 inputs (0, 180, 90, and 270 degrees) and 4outputs (90, 180, 90, and 270 degrees). However, only the 0 and 180degree inputs are utilized to receive the LO signal 313. The other 2inputs, 90 and 270 degrees, can be either grounded or left open. TheInventors have found that grounding the 90 and 270 degree input providesbetter noise performance.

As discussed herein, four VCOs (318 a-d) and two polyphase circuits (310and 312) are used to cover the desired LO frequency range from 950 MHzto 2150 MHz. However, more or less VCOs and/or polyphase circuits couldbe utilized to cover the LO frequency range depending on the desiredperformance specifications, as will be understood by those skilled inthe relevant arts. These other configurations that utilize a differentnumber of VCOs and/or polyphase circuits are within the scope and spiritof the present invention.

Furthermore, the VCOs 318 are configured to operate over a frequencyrange of 950-2150 MHz. However, the present invention is not limited tooperating over this frequency range. Other frequency ranges could bechosen as will be understood by those skilled in the arts based on thediscussion given herein. These other frequency ranges are within thescope and spirit of the present invention.

Quadrature Mixers

As stated above, IQ mixers 334 down-convert the differential RF signal305 directly to baseband by mixing with the differential RF signal 305with the corrected LO signal 333. FIG. 6 further illustrates the I mixer334 a and the Q mixer 334 b as a modified differential Gilbert cellmixer comprised of multiple FET transistors and variable current sources602 and 604. The mixer 334 in FIG. 6 is described generically withoutregard to I and Q signals, as I/Q distinction only results in a 90degree phase change between the relevant signals.

Referring to FIG. 6, an RF transconductance circuit 616 receives thedifferential RF signal 305 and converts the differential signal 305 to adifferential current 622. More specifically, a FET 618 receives apositive voltage component 305 ⁺ of the differential RF signal 305 andconverts it to an RF current 622 ⁺. Similarly, a FET 620 receives anegative voltage component 305 ⁻ and converts it to an RF current 622 ⁻.(Herein, the individual positive and negative components fordifferential signals are identified using “+” and “−” designations afterthe reference numbers.)

A switching circuit 606 receives the RF current 622 and commutates theRF current 622 according to the LO signal 333, to produce a basebandoutput 335. More specifically, FETs 608 and 610 commutate the RF currentI₁ 622 ⁺ according to a positive component 333 ⁺ and a negativecomponent 333 ⁻, respectively, of the LO signal 333. As result, thecurrent RF current I₁ 622 ⁺ is switched between the outputs 335 ⁺ and335 ⁻ at the clock rate of the LO signal 333. Similarly, FETs 612 and614 commutate the RF current I₂ 622 ⁻ according to the positivecomponent 333 ⁺ and the positive component 333 ⁻, respectively, of theLO signal 333. As result, the current RF current I₁ 622 ⁻ is switchedbetween the outputs 335 ⁺ and 335 ⁻ at the clock rate of the LO signal333. The sampling of the RF signal 622 at the rate of the LO signal 333produces the signal mixing action and frequency translation. Since thefrequency of the LO signal 333 is centered on a selected channel of theRF signal 305, the mixing action of the FETs 608-614 produces sum anddifference frequencies at the outputs 335, where the differencefrequency is at approximately baseband, depending on the accuracy of theLO signal 333.

In order to minimize the flicker noise, the FETs 618 and 620 should beoperated at relatively high current flow in I₁ and I₂. Ignoring (for themoment) the variable DC current sources 602 and 604, the currents I₁ 622⁺ and I₂ 622 ⁻ flow through the respective FETs 608-614 in the switchingcircuit 606, which are controlled by the LO signal 333. If the currentI₁ 62 ⁺ and I₂ 622 ⁻ are sufficiently large, then the LO signal 333would also need to be large to switch FETs 608-614 completelyon-and-off. Mixer gain would decrease and flicker noise would increaseif the FET 608-614 do not turn completely “on” or completely “off”because of inadequate LO drive. However, there are limitations to thesignal amplitude that is available from the LO correction circuit 329.Furthermore, the greater the current that flows through the FETs608-614, the larger the devices need to be to handle this current.Larger FET devices have larger parasitic capacitance that reducesoverall signal bandwidth. Thus, there is a trade-off between the desireto maximize the current flow 622 to minimize flicker noise in the FETs608-614 and the desired to limit the LO drive requirement of the LO 333.

The variable current DC sources 602 and 604 are added to the mixer 334to address the mentioned tradeoff between flicker noise and LO drive.The variable current sources add DC current to the signals I₁ 622 ⁺ andI₂ 622 ⁻. More specifically, the current generator 602 generates a DCcurrent 624 a that adds to the current I₁ 622 ⁺. Likewise, the currentgenerator 604 generates a DC current 624 b that adds to the current I₁622 ⁻. The currents 624 a and 624 b do not flow through the mixing FETs608-612, and therefore are not involved in the mixing process performedby the FETs 608-612. As a result, the currents 622 ⁺ and 622 ⁻ areincreased without increasing the amplitude requirements on the LO drive333, and without increasing the size requirements of the FET devices608-612. The DC current sources 602 and 604 are set to providepredetermined portions of the total current 622 ⁺ and 622 ⁻,respectively, and can be adjusted as desired. In other words, the DCcurrent sources 602 and 604 can be adjusted to minimize the flickernoise of the FETs 608-614, as desired.

The DC currents 624 a and 624 b can be referred to as “bleedercurrents”, because they bleed-off some of the current requirements thatflow through the mixing FETs 608-612.

FIG. 7 further illustrates the mixer 334. In FIG. 7, the variable DCcurrent source 602 is implemented as cascode-connected FETs 704 and 706to generate the bleeder current 624 a. The FETs 704, 706, 708, and 710are biased at their respective gates to provide the desired bleedercurrents 624 a and 624 b to reduce flicker noise. The variable DCcurrent source is implemented as cascode-connected FETs 708 and 710 togenerate the bleeder current 624 b. Bias for the variable DC currentsources 602 and 604 are connected to the nodes 714 and 716, whichprovides the proper gate voltages to produce the desired bleedercurrents 624 a and 624 b. Furthermore, FIG. 7 further illustrates themixer 334 having capacitors 702, 703, 711, 712 that provide some lowpassfiltering for the output signal 335. The devices 718 and 720 are spareor dummy devices.

Tunable Baseband Filters

As stated above, the tuneable baseband filters 344 a,b lowpass-filterthe I and Q baseband signals 343 a,b to remove unwanted frequencies, soto produce baseband signals 345 a,b. FIG. 8 further illustrates thetunable baseband filter 344 a or 334 b according to embodiments of thepresent invention. Referring to FIG. 8, the tunable baseband filter 344includes a group of integrators 804 a-e that are series connected with agroup of corresponding summers 802 a-e. As illustrated, the output ofthe nth integrator 804 is fed back to the (n−1)^(th) associated summer802. For example, the output of the integrator 804 b is fed back to theinput of the summer 802 a, and the output of the integrator 804 c is fedback to input of the summer 802 b.

The tuneable lowpass filter 344 simulates the operation of aconventional Butterworth lowpass filter 900 that is shown in FIG. 9. Theintegrators 804 simulate the corresponding elements of the lowpassfilter Butterworth filter 900, where the Butterworth elements areindicated over the integrator 804 in FIG. 8. For example, the integrator804 a simulates the electrical response of the 1/RC₁ in the Butterworthfilter 900, and the integrator 804 b simulates the electrical responseof the R/L₂ in the Butterworth filter 900.

The Butterworth filter 900, and all Butterworth filters, have a knownfrequency response and cutoff frequency, which can be used as a designmethodology for the lowpass filter 800. For a given filter specification(e.g. cutoff frequency, in-band ripple, etc.), the circuit elements(i.e. R, L, C values) of the Butterworth filter 900 can be calculatedusing known techniques. Afterwhich, the active integrators 804 areconfigured to simulate the circuit elements in the Butterworth filter900. The specific configuration of the integrators 804 in FIG. 8 isoften referred to as a “Leapfrog” synthesis or configuration. Otherconfigurations could be used to simulate a passive Butterworth withactive filter elements, and are known to those skilled in the arts.However, it has been found that the leapfrog configuration provides thelowest sensitivity to circuit component variation, and provides goodnoise performance. Furthermore, other types of filters, other than aButterworth, could be used depending on the application.

An exemplary stage 806 of the filter 344 includes the summer 802 c andthe integrator 804 c. FIG. 10 further describes the exemplary stage 806as an active op amp RC filter having resistors 1002 a-d, capacitors 1004a,b, and an operational amplifier 1006, all of which are configured tobe differential. The operational amplifier 1006 provides the integrationfunction for the integrator 804 over the filter response that isdetermined by the capacitors 1004 and the resistor 1002 a-d. Forexample, the resistors 1002 provide the resistance associated withresistor 1/RC₃ in the integrator 804 c, and the capacitors 1004 a,bprovide the capacitance associated with the integrator 804 c. Thedifferential voltages V₂ (output of integrator 804 b) and V₄ (output ofintegrator 804 d) are wire-ored together at the input of the operationalamplifier 1006 to construct the summer 802 c.

In one embodiment of the invention, the capacitors 1004 are realizedusing metal-oxide semiconductor (MOS) devices that are configured ascapacitors (MOSCAPs). Alternatively, the capacitors 1004 are metal-metalcapacitors. MOSCAPs produce a variable capacitance depending on thevoltage that is placed across the MOSCAP. For example, FIG. 11illustrates stages 806 and 808 having MOSCAPs 1104 to implement thecapacitors 1004. The capacitance provide by each MOSCAP 1104 isdetermined by the bias voltages V_(B1) and V_(B2) that are placed acrossthe MOSCAP, where V_(B2) is generally greater than V_(B1). For example,V_(B2) may be 2.3V and V_(B1) may be 1.3V, to produce a desiredcapacitance. Since the filter stages 806 and 808 are connectedin-series, the level shifters 1106 a,b are inserted between the stages806 and 808 to level shift bias voltage V_(B2) from stage 806 to V_(B1)for the stage 808. Without the level shifters 1106, then V_(B2) from thegate of MOSCAP 1104 (stage 806) would bleed over to the source of theMOSCAP 1104 c, eliminating the voltage bias across MOSCAP 1104 c. Thelevel shifters 1106 create a DC offset (from V_(B2) to V_(B1)) betweenthe stages 806 and 808, permitting the use of the MOSCAPs 1104 for thecapacitors 1004. As will be apparent to those skilled in arts, the levelshifters 1104 can be inserted between all the stages of the filter 344,to permit the MOSCAPs 1104 to be used in all the stages of the filter344.

FIG. 12 illustrates one embodiment of the level shifter 1106 that isknown as a source follower circuit that drops the voltage from V_(B2) toV_(B1) based the voltage drop through the resistor 1204. Preferably, theg_(m) of the FETs 1202, 1206 is sufficiently large to minimize signalnon-linearity and to drive a low resistor 1204. An advantage of thelevel shifter in FIG. 12 is that the op amp 1006 only needs to drive acapacitance load, which relaxes the design requirement on the op amp1006. Furthermore, there is no headroom issue with the level shifter1106, so the size of the current source 1206 can be made small resultingin a small parasitic capacitance, and therefore good overall bandwidth.

As discussed herein, it is desirable to tune the cutoff frequency of thebaseband filter 344 to accommodate changing baseband bandwidthrequirements for various service providers. For example, in embodimentsof the invention, it is desirable to tune the lowpass cutoff frequencyfrom 2 MHz to 36 MHz to provide for varying satellite service providerrequirements. Referring to FIG. 10, this can be accomplished by tuningeither the resistors 1002 or the capacitors 1004. Capacitor tuning istypically performed with a bank of parallel capacitors that areincrementally switched-in or out by operating corresponding switchesthat are in series with the capacitors. However, when the preferredMOSCAP are utilized, tuning capacitance is not desirable because a bigMOS switch is needed to minimize the resistance, resulting in a largeparasitic capacitance that reduces bandwidth and provides poor tuningaccuracy. As a result, it is preferable to tune the resistors 1002 toaccomplish tuning of the lowpass cutoff frequency.

FIG. 13 further illustrates an example tunable resistor 1002 for thepurpose of tuning the cutoff frequency of the lowpass filter 344.Referring to FIG. 13, the tunable resistor 1002 includes a plurality ofresistors 1302 that are connected in parallel, where each resistor 1302includes a series-connected switch that can be a MOSFET switch. Thecutoff frequency of the lowpass filter 344 is dependent on 1/RC.Therefore, the cutoff frequency can be adjusted by incrementally addingor moving the resistors 1302 to or from the resistor 1002. Inembodiments of the invention, the resistors 1302 are binary weighted. Inother words, the value of the resistors 1302 have a binary relationshipas illustrated in FIG. 13. For example, the resistor 1302 a can have avalue of R, and the resistors 1302 b-1302 f can have the values R/2,R/4, R/8, R/16, and R/32, respectively. The binary relationship canproduce frequency tuning in pre-defined frequency steps. For example,the cutoff frequency can be tuned in 1 MHz frequency steps using a 6-bitbinary weighted resistors 1302, for a 5th order Butterworth havingcapacitive elements C1=3.93pF, C2=10.85 pF, C3=12.73 pF. The cutofffrequency can be tuned in 1 MHz steps, where R is determined accordingto the following equation:R=(25 Kohm)/(1 MHz*f _(C))  (1)

The ability to tune the cutoff frequency of the lowpass filter 344 inpredefined frequency steps allows the same tuner 306 to be used bymultiple service providers that may have different baseband bandwidthrequirements. The tuner 306 can be mass produced without regard for thebaseband bandwidth requirement of the ultimate service provider. Once aservice provider is identified, the cutoff frequency of the lowpassfilter 344 can be adjusted in predefined frequency steps byincrementally adding or subtracting the resistors 1304 by closing theappropriate switches.

Furthermore, the decision to tune the cutoff frequency by adjusting theresistors in the lowpass filter 334, means that the capacitors 1004 canbe fixed. This is advantageous because tuning capacitors in a MOSFETdevice requires significantly larger substrate area than tuningresistors. In other words, adjusting the resistors 1002, instead of thecapacitors 1004, allows the tunable lowpass filter 344 to be integratedon the same substrate with the rest of the components of the tuner 306,instead of being placed off-chip.

In alternative embodiments of the invention, the tunable resistors 1002are variable “analog” resistors that are continuously variable.

DC Offset Compensation

As discussed herein, the tuner 306 is preferably configured to bedifferential to optimize noise performance. Differential signals have apositive signal component and a negative signal component. Componentmismatches can produce a DC voltage offset to develop between thepositive signal component and the negative signal component. DC offsetvoltages are undesirable because they can easily saturate a differentialamplifier and other components in a differential system.

AC coupling is one known technique for addressing DC offset. AC couplinguses series capacitors to block any DC voltage buildup, thereby removingany DC offset. However, since the capacitors are series coupled, theymust be sufficiently large to pass the desired AC signal. At basebandfrequencies, the capacitors would be extremely large, preventing theintegration of these capacitors on-chip. Therefore, the presentinvention utilizes a DC compensation circuits 348 to sense the DC offsetvoltage at the tuner output 114. The DC compensation circuits 348 removethe DC offsets from the respective I and Q baseband signals 114 a and114 b using a subtractive feedback technique, without the need forseries capacitors.

FIG. 14 further illustrates the DC compensation circuit 348 for eitherthe I channel or the Q channel, without regard for the in-phase orquadrature signal status. The DC compensation circuit includes the VGA338, the baseband filter 344, the amplifier 346, and the DC servo 342.The DC Servo 342 provides negative fed back path for the DC offset andcancels the DC offset at the output of the VGA 338.

The DC servo 342 includes two transconductance amplifiers 1406,1410, anda lowpass filter 1408. The amplifier 1410 senses the positive component114 ⁺ and the negative component 114 ⁻ of the output signal 114, andamplifies any voltage difference between the positive component 114 ⁺and the negative component 114 ⁻. The amplifier 1408 is a differentialtransconductance amplifier that converts any voltage difference to adifferential current 1409. In other words, the greater the voltagedifference, the larger the differential current output 1409 of theamplifier 1408. The lowpass filter 1408 removes all the high frequencycontent from the differential current 1409 and passes only the lowfrequency content, which contains the DC offset voltage 1407. Inembodiments of the invention, the cutoff frequency of the lowpass filter1408 is set to 100 Hz (or below), so that only the DC offset voltage1407 is passed to the amplifier 1406. The amplifier 1406 is atransconductance amplifier that converts the DC offset voltage 1407 to acorresponding differential DC offset current 1405 that is sent to theVGA 338.

The VGA 338 is two stage amplifier having a first amplifier 1402 and asecond amplifier 1404. The first amplifier 1402 is a transconductanceamplifier that converts the differential baseband signal 337 from adifferential voltage to a differential current 1403. The output of theamplifier 1406 in the DC servo 342 is combined with the output of theamplifier 1402 at the nodes 1412 and 1414. More specifically, thepositive output of the DC servo amplifier 1406 is connected to thenegative output of the amplifier 1402 at the node 1414. Likewise, thenegative output of the DC servo amplifier 1406 is connected to thepositive output of the amplifier 1402. Therefore, the DC offset current1405 is combined 180 degrees out-of-phase with the differential basebandcurrent 1402 at the nodes 1412 and 1414, canceling any DC offset in thebaseband current 1402. The resistors 1416 and 1418 convert thedifferential current at the nodes 1412 and 1414 to a differentialvoltage 1420. The differential 1420 corrects any DC offset introduced byamplifier 1402 and all subsequent stages 1404, 344, and 346, to minimizeor eliminate the offset at the outputs 114 ⁺ and 114 ⁻. The differentialvoltage 1420 is received at the variable amplifier 1404 that iscontrolled by the AGC signal 350. The variable amplifier 1404 variablyamplifies the differential voltage 1420 according to the AGC signal 350,to generate the VGA output signal 343 that is free of DC voltage offset.

An advantage of the DC compensation circuit 348 is that the DC offsetsignal 1405 is fed back to the input of the amplifier 1404 of the VGA338, instead of to the output of the amplifier 1404 in the VGA 338. Thisconfiguration has a better noise performance than the combining the DCoffset signal 1405 at the output of the VGA 338.

As stated above, the lowpass filter removes all the higher frequencycontent from the differential current 1409 and passes only the DC offsetvoltage 1407. It is important reject the higher frequency content in thedifferential current 1409 so that it does not cancel the intendedbaseband information in the baseband signal 1403. In other words, it isimportant that only the DC offset gets canceled when combining signals1403 and 1405. In embodiments of the invention, the lowpass filter 1408is capacitor 1502 that is shown in FIG. 15. The capacitor 1502 isconnected across the differential output of the DC servo amplifier 1410and is large enough to short out the higher frequency content in thedifferential current 1409, with the exception of DC offset voltage. Inembodiments of the invention, the capacitor sufficiently large to shortout all signals above 100 Hz, and thereby substantially passes only theDC offset voltage to the amplifier 1406. Furthermore, in embodiments ofthe invention, the capacitor 1502 is located off-chip, due to its size.Alternatively, if a smaller capacitor can be used, the capacitor 1502can be located on-chip.

The flowchart 1600 summarizes the operation of the DC compensationcircuit 338 in removing DC offset from an input differential basebandsignal, according to embodiments of the present invention.

In step 1602, the input differential baseband signal is received by theDC compensation circuit and converted to an input differential basebandcurrent. For example, the transconductance amplifier 1402 receives thedifferential baseband signal 337 and converts it to a differentialbaseband current 1403. At this point, any DC offset that is present isstill in the differential baseband current 1403.

In step 1604, the differential output voltage of the DC compensationcircuit is sensed and converted to a differential current. For example,the transconductance amplifier 1410 converts the differential outputvoltage 114 to a differential current 1409.

In step 1606, the differential output current is lowpass filtered tosubstantially pass only the DC offset information and to reject thebaseband signal information, producing a differential signalrepresentative of the DC offset. For example, the lowpass filter 1408filters the differential output current 1409 to produce a differentialvoltage 1407 that substantially contains only the DC offset voltage.

In step 1608, the DC offset voltage from the lowpass filter is amplifiedand converted to a differential DC offset current that represents the DCoffset voltage. For example, the transconductance amplifier 1406converts the DC offset voltage 1407 to a differential current 1405.

In step 1610, the differential DC offset current is inverted so that itis 180 degrees out-of-phase with the differential input basebandcurrent. For example, the output of the transconductance amplifier 1406is connected to the output of the transconductance amplifier 1402 sothat the positive and negative terminals are reversed at the nodes 1412and 1414.

In step 1612, the inverted DC offset current is combined with the inputbaseband current so that the DC offset at the output is canceled. Forexample, the DC offset current 1405 is combined with the input basebandcurrent 1403 at the nodes 1412 and 1414, canceling any DC offset goinginto the output amplifier 1404 of the VGA 338, and canceling any DCoffset at the output of the VGA 338.

Baseband Filter Tuning

As stated above, the baseband filters 344 are tunable to accommodate thevarying bandwidth requirements of different service providers. Forexample, in embodiments of the invention, it is desirable to tune thecutoff frequency of the baseband filters from 2-36 MHz using the I²Ccontrol 354. In embodiments, it is desirable to tune the cutofffrequency in 1 MHz increments. As described herein, and referring toFIG. 10, the filter tuning is performed by varying the resistor values1002 in exemplary filter stage 806 to adjust the cutoff frequency of thetuner 306. FIG. 17 and the discussion that follows further describestuning the cutoff frequency of the baseband filter 344 by tuning thevariable resistors 1002. FIG. 17 is depicted as single-ended instead ofdifferential for ease of illustration, but the tuning proceduredescribed in relation to FIG. 17 applies equally well to differentialconfigurations as will be understood by those skilled in the relevantarts.

FIG. 17 illustrates an active low-pass filter and compensation circuitfor achieving accurate filtering on an integrated circuit. In order toovercome manufacturing process variations and errors introduced bytemperature variations, an active low-pass filter 1702 has been combinedwith a compensation circuit 1704 in order to accurately adjust thecorner frequency of the low-pass filter 1702. (Low pass filter 1702 is asingle ended version of baseband filter stage 806 in FIG. 10). Ingeneral, the compensation circuit 1704 generates a control signal 1706to adjust two variable resistors (1716 and 1720). Variable resistor 1716in the compensation circuit 1704 and variable resistor 1720 in theactive low-pass filter 1702 are substantially identical and are adjustedby control signal 1706 as a function of the equivalent resistance of aswitched-capacitor 1708, and the V_(adj)/V_(ref) ratio that determinesthe corner frequency of the lowpass filter.

Specifically, compensation circuit 1704 comprises switched capacitor1708, an amplifier 1710, a comparator 1712, a successive approximationregister (SAR) 1714 and a variable resistor 1716. An adjustable voltage(V_(ADJ)) is applied to an input of the switched-capacitor 1708. Anoutput of switched-capacitor 1708 is coupled to an inverting input ofamplifier 1710. A non-inverting input of amplifier 1710 is coupled toground. An output of amplifier 1710 is coupled to an inverting input ofa comparator 1712.

A reference voltage (V_(REF)) is coupled to a non-inverting input ofcomparator 1712. An output of comparator 1712 is coupled to an input ofA/D converter 1714. The A/D converter 1714 produces the control signal1706, which is described in further detail below. The variable, orotherwise adjustable resistor (R_(ADJ)) 1716 is coupled between theinverting input of amplifier 1710 and its output (which is also theinverting input of comparator 1712). Control signal 1706 is also coupledto R_(ADJ) 1716 to change its resistance value.

The active low-pass filter (LPF) 1702 comprises a variable resistor1720, a capacitor 1722 and an amplifier 1724. A signal to be filtered isapplied to a first node label V_(IN), which is coupled to resistor 1720.Resistor 1720 also coupled to the inverting input of amplifier 1724. Anon-inverting input of amplifier 1724 is coupled to ground. Capacitor1722 is coupled across the inverting input of amplifier 1724 and itsoutput node, which is labeled as V_(OUT). Variable resistor 1720 alsoreceives control signal 1706 to change its resistance value.

Operation of the compensation circuit 1704 in FIG. 17 will be describednext. To illustrate the operation of compensation circuit 1704, considera case in which voltages V_(ADJ) and V_(REF) are kept constant. Also,for this explanation, assume comparator 1712 and SAR 1714 simplycomprise a amplifier 1730 that produces the control signal 1706 toadjust resistor 1716. In the simple case, the amplifier 1730 willproduce a control signal 1706 to adjust resistor 1716 to match the valueof resistor 1708 until the output voltage of amplifier 1710 is equal toV_(REF). Thus, once the voltage levels at the input of amplifier 1730are the same, control signal 1706 will no longer change the resistanceof resistor 1716.

In order to establish an accurate corner frequency for the activelow-pass filter 1702, the product of the resistance value for resistor1720 and the capacitance value for capacitor 1722 must be accurate.Since a stable capacitance value can be achieved using existingsemiconductor manufacturing techniques, an initial capacitance value forcapacitor 1722 can be determined. However, because of processingvariations and temperature variations, the exact capacitance of thecapacitor can vary from chip-to-chip. Because the capacitance would varyfrom chip-to-chip the corner frequency will also vary even though anaccurate fixed resistance value for th resistor 1720 is provided, asdescribed above.

The exact corner frequency, however, can be achieved by varying theresistance of resistor 1720 to an exact resistance value equal to1/[2πf_(C)C], where f_(C) is the corner frequency of th low-pass filter.This can be achieved using a switch-capacitor circuit for R_(SC) 1708.

In embodiments, a capacitance value and switching frequency value areselected for switching-capacitor 1708 in order to achieve the exactdesired resistance for the active low-pass filter 1702. In operation,since the ideal amplifier 1730 produces control signal 1706 so as tocause the resistance of variable resistor 1716 to match the resistanceof switch capacitor 1708, control signal 1706 is also supplied tovariable resistor 1720. Thus, by achieving a desired equivalentresistance at switched capacitor 1708, the compensation circuit 1704,via amplifier 1730, will produce a control signal 1706 so as to causeresistor 1720 of the active low pass filter 1702 to produce a resistancevalue for resistor 1720 equal to the affective resistance ofswitch-capacitor 1708 equal to 1/[f_(CLK)*C_(SC)], where F_(CLK) is theswitching frequency and C_(CS) is the capacitance value of C_(S) in theFIG. 18. Equating this resistance value to 1/[2πf_(C)C] in order to getthe desired accurate corner frequency of the low pass filter will bedescribed below. In summary, the resistance of variable resistors 1716and 1720 is adjusted via the control signal 1706 until the desired valuefor the LPF corner frequency is achieved.

According to one embodiment of the present invention, adjusting f_(CLK)of the switched capacitor 1708 will change its resistance. Tocompensate, amplifier 1730 adjusts control signal 1706 to change thevalue of resistor 1720, thereby changing the LPF corner frequency of theactive low-pass filter 1702.

According to another embodiment of the present invention, control signal1706 is changed by adjusting a ratio “K” of voltages V_(ADJ) and V_(REF)(i.e., K=V_(ADJ)/V_(REF)), while F_(CLK) remains constant:

$\begin{matrix}{R = \frac{1}{C_{SC} \cdot f_{CLK} \cdot K}} & (2)\end{matrix}$Changing the ratio “K” causes the differential voltage at the input ofamplifier 1730 to change. To compensate, the amplifier 1730 changescontrol signal 1706 so as to vary the resistance of adjustable resistorR_(ADJ) 1716, thereby causing the voltage at its inverting input toagain match the voltage at its non-inverting input. At the same timecontrol signal 1706 adjusts the resistance of resistor 1716 tocompensate for the changed voltage ratio, control signal 1706 alsochanges the resistance of resistor 1720 thereby changing the cornerfrequency of the active low-pass filter 1702. In order to tune thecorner frequency of the low-pass filter from 2 MHz to 36 MHz, K isvaried from 1 to 18 respectively in this design. However, in order tomake the design insensitive to errors produced by the non-idealities ofthe switch capacitor circuit 1708 and the operational amplifier 1710,high values of K are desired. By dividing f_(CLK) for lower-half cornerfrequencies, K is circulated from 8 to 18 instead of changing from 1 to18. This technique improves the circuit sensitivity for cornerfrequencies from 2 MHz to 15 MHz. The sensitivity is further improved byreducing the offset voltage of the operational amplifier 1710 and thecomparator 1712 by employing an offset-cancellation scheme in thecomparator 1712.

Also, according to this latter embodiment of the present invention, theratio of voltages V_(ADJ) and V_(REF) can be changed by changing eitherV_(ADJ) or V_(REF), or both V_(ADJ) and V_(REF). Preferably, V_(REF) canbe set to a constant reference voltage, while voltage V_(ADJ) isadjusted so as to change the corner frequency of the active low-passfilter 1702. The voltages V_(ADJ) and V_(REF) can be implemented using aresistor ladder with variable tap points. Other voltage sources can beused to provide V_(ADJ) and V_(REF), as would become apparent to aperson skilled in the relevant art.

The switch capacitor C_(SC) in FIG. 18 is implemented as a NMOS in NWELLcapacitor, the same way for the C 1722 in the low-pass filter. In oneembodiment, F_(CLK) is equal to 16 MHz, and the value of Csc is scaledto be π*C/4 in order for R in Equation 2 above to be equal to1/(2πf_(C)C). As a result, the desired accurate corner frequency of thelow-pass filter will be established.

Variable resistors 1716 and 1720 can be implemented in a variety ofways. Each can comprise a bank of selectable resistors, for example.Other equivalent resistor networks will become apparent to a personskilled in the relevant art.

Control signal 1706 can be a digital signal so as to select one or moreof the individual resistors in each respective resistor bank. In orderto produce a digital control signal 1706, the analog-to-digitalconverter 1730 can comprise a comparator 1712 coupled to a SAR 1714.Other equivalent circuit to implement the functionality of amplifier1730 for generating control signal 1706 will become apparent to a personskilled in the relevant art.

The switched capacitor 1708 is further illustrated in FIG. 18. Aswitched capacitor is based on the realization that a capacitor switchedbetween two circuit nodes at a sufficiently high rate is equivalent to aresistor connecting these two nodes. Specifically, the two switches S₁and S₂ of FIG. 18 are driven by a non-overlapping, two-phase clock,F_(CLK). During clock phase φ₁ capacitor C_(S) charges up to a voltageat node 1802 by closing S₁. Then, during a second clock phase φ₂,capacitor C_(S) is connected to the output node 1804 by closing S₂, andthe capacitor C_(S) is forced to discharge, transferring its previouscharge to the output node 1804. Thus, if f_(CLK) is much higher than thefrequency of the voltage wave forms of V_(IN), then the switchingprocess can be taken to be essentially continuous, and aswitched-capacitor can then be modeled as an equivalent resistance asshown in the equation below:

$\begin{matrix}{R_{eq} = \frac{1}{C \cdot f_{CLK}}} & (3)\end{matrix}$

For illustration purposes, FIG. 18 includes an inverter 1806 thatinverts f_(CLK) to generate the opposite phase clock φ₂. Any personskilled in the relevant art will recognize that the non-overlappingclocks φ₁ and φ₂ can be produced in many ways. Moreover, switches S₁ andS₂ can be implemented with transistors (for example, metal oxidesemiconductor field affect transistors (MOSFETs), or the like).Additionally, various means are commercially available for generatingthe clock frequency.

The tuning of the low pass filter 344 is further described in co-pendingpatent application, entitled, “Low Pass Filter Corner Frequency TuningCircuit and Method”, Ser. No. 09/995,795, filed herewith, and which isincorporated by reference herein in its entirety.

Low Noise Amplifier

As discussed herein, the LNA 304 receives the differential RF signal 303from the off-chip balun 302. The LNA 304 variably amplifies thedifferential RF signal 303 according to an RF AGC control 352, toproduce a differential RF signal 305. The differential RF signal 305 isforwarded to the RF input of the IQ mixers 334 for down-conversion.

FIG. 19 further illustrates the LNA 304 according to embodiments of theinvention. The LNA 304 includes a combination attenuator and matchingcircuit 1902 (herein called attenuator 1902 for convenience), a controlcircuit 1930, a gain block 1916, and an output matching circuit 1928.The gain block 1916 includes a high gain amplifier 1918 and a low gainamplifier 1922. The differential RF signal 303 is coupled directly tothe high gain amplifier 1918. The attenuator 1902 also receives thedifferential RF signal 303 and generates an attenuated differential RFsignal 1921 that is coupled to the low gain amplifier 1922. The highgain amplifier 1918 variably amplifies the differential RF signal 303,and the low gain amplifier 1922 variably amplifies the attenuateddifferential RF signal 1921. The outputs of the high gain amplifier 1918and the low gain amplifier 1922 are combined in a summer 1920 to producean amplified output 1927. The output matching circuit 1928 provides animpedance match for the output signal 1927 to the IQ mixers 334, toproduce the LNA output signal 305 that is sent to the IQ mixers 334.

The high gain amplifier 1918 and the low gain amplifier 1922 include aplurality of amplifiers that are configured in-parallel and arecontrolled in groups by AGC control signals 1924 from the controlcircuit 1930. The AGC controls signals 1924 are generated by DCamplifiers 1923 based on the master AGC control signal RF AGC 352 (FIG.3). The control circuit 1930 also generates bias currents 1926 for theamplifiers in the high and low gain amplifiers 1918 and 1922, based onthe master AGC control signal RF AGC 352. The structure and operation ofthe components of the LNA 304 are described further below.

The attenuator 1902 includes resistors 1906 a and 1906 b, resistors 1910a and 1910 b, and tapped inductors 1908 a and 1908 b. The attenuateddifferential RF signal 1921 is tapped-off the center of the tappedinductors 1908 a and 1908 b. The tap point on the inductors 1908 ischosen to provide a desired attenuation for the attenuated signal 1921.In embodiments of the invention, the inductors 1908 are tapped so as toprovide 12 dB of attenuation. The amplifiers 1918 and 1922 areimplemented using MOSFETs that have a capacitive input. The tappedinductors 1908 help match the input impedance looking in to theattenuator 1902 to a real impedance over the frequency band of interest.The remaining component values of the attenuator 1902 are determined toprovide the desired attenuation and to provide a good impedance match tothe amplifiers 1918 and 1922 over the frequency band-of-interest, (950MHz to 2150 MHz). More specifically, the resistors 1906 and 1910contribute to a flat amplified match over the desired frequency band.Therefore, the combination of the tapped inductors 1908 and theresistors 1910 and 1906 provide a flat and smooth impedance match overfrequency, and well as providing the desired signal attenuation. Inembodiments of the invention, the resistors 1906 a,b are chosen to be23.1 ohms and resistor 1910 a,b are chosen to be 8.7 ohms. By addingsome minimal off-chip matching elements, this provides 75 ohm inputimpedance over the desired frequency band of 950-2150 MHz at the outputof the balun 302.

The attenuator 1902 also includes a node 1911 between resistors 1910 aand 1910 b. This node 1911 can be grounded through a capacitor 1905 to acenter tap 1904 of the attenuator 1902. Alternatively, the center tap1904 can be left ungrounded and used for a single-ended input.

The tapped inductors 1908 a,b provide a DC path to the FETs in the highgain amplifier 1918 and the low gain amplifier 1922. Therefore, a biasvoltage can be placed at the node 1914, to bias the FETs in the highgain amplifier 1918 and the low gain amplifier 1922. The capacitor 1905DC blocks the bias voltage at the node 1914 from the ground at thecenter tap 1904. Furthermore, the inputs 303 are also DC blocked. Sincea FET gate draws practically zero current, there is no voltage dropacross the resistors 1912, 1910, or 1906. Furthermore, the resistor 1912is large (greater than 10K ohms) so as not to load AC signal.

FIG. 20 further illustrates the gain block 1916 and the configuration ofthe high gain amplifier 1918, the low gain amplifier 1922, and thesummer 1920. The high-gain amplifier 1918 includes a set of 3 binaryweighted gain stages identified as ½ gain stage, ¼ gain stage, ⅛ gainstage_a, which amplify the unattenuated differential RF signal 303. Thelow gain amplifier 1922 includes another set of 3 binary weighted gainstage identified as ⅛ gain stage_b, 1/16 gain stage, and 1/32 gainstage, which amplify the attenuated differential RF signal 1921. Eachgain stage contains one or more identical parallel-connected unit gainamplifiers 2002 having outputs that are wire-ored together, thus formingthe summer 1920. In other words, all the outputs of the high gain stage1918 are wired-ored with all the outputs of the low gain stage 1922. Thenumber of unit amplifiers in each gain stage varies depending on therelative gain assigned to each stage. For example, the ½ gain stageinclude 16 out of 35 total amplifiers 2002 that are shown in FIG. 19.The ¼ gain stage includes 8 out of 35 total amplifiers 2002, and so on.Since the outputs of the binary weighted stages are summed, the ½ gainstage provides approximately ½ of the gain of the LNA 304, the ¼ gainstage provides approximately ¼ of the gain, and so on. As will beapparent, the invention is not limited to the number and grouping ofamplifiers 2002 that are shown in FIG. 19. Other configurations andamplifier quantities could be utilized as will be understood by thoseskilled in the arts. These other configurations are within the scope andspirit of the present invention.

The AGC signals 1924 are differential and adjust the gain of thecorresponding binary weighted gain stage by adjusting the gain of theunit gain amplifiers 2002. For example, the AGC 1924 _(—)½ controls theunit gain amplifiers 2002-1 to 2002-16, the AGC 1924 _(—)¼ controls theunit gain amplifiers 2002-17 to 2002-24, and so on. Likewise, the biascontrol signals 1926 supply and adjust the current bias for the unitgain amplifiers 2002 in the corresponding binary gain stages. Forexample, the bias control signal 1926 _(—)½ controls the unit gainamplifiers 2002-1 to 2002-16 in the ½ gain stage. Likewise, the biascontrol signal 1926 _(—)¼ controls the unit gain amplifiers 2002-17 to2002-24 in the ¼ gain stage, and so on.

FIG. 21 further illustrates an example unit gain amplifier 2002.Referring to FIG. 21, the unit gain amplifier 2002 includes a variablecurrent source 2122, a differential pair 2116 having FETs 2118 and 2120,an output steering circuit 2102 having FETs 2104-2110, and bleederresistors 2112 and 2114. The differential pair 2116 receives the RFdifferential signal 303 or 1921 depending on whether the unit gainamplifier 2002 is in the high gain amplifier 1918 or the low gainamplifier 1922. The differential pair 2116 amplifies the RF differentialsignal 303 (1921) to produce a differential output current 2117. Theoutput steering circuit 2102 receives the differential output current2117 and steers the current 2117 to the amplifier output nodes 2101 orto the analog power supply av_(dd), wherein the current steering iscontrolled by the corresponding agc signal 1924. More specifically, theagc signal 1924 controls the gates of the FETs 2104-2110 so as to steerthe differential output current 2117 to either the output 2101 or to theoutput av_(dd). The bleeder resistors 2112 and 2114 bleed-off some ofthe current from the differential pair 2116 to improve overall circuitheadroom.

During operation, the differential agc signal 1924 is increased whenmore gain is required, so that more of the differential output current2117 is steered to the output 2101 instead of the analog power supplyav_(dd). Likewise, the agc signal 1924 is reduced when less gain isrequired, so that more of the differential output current 2117 issteered to the analog power supply av_(dd), instead of to the outputnodes 2101. At some point during gain reduction (called the 0-gainthreshold), substantially all of the current 2117 is steered to thepower supply av_(dd). When this occurs, the unit gain amplifier 2002 isno longer providing any gain to the LNA 304.

As indicated in FIG. 20, all the unit gain amplifiers 2002 in a binaryweighted gain stage are controlled by the same AGC signal 1924. Forexample, the unit gain amplifiers 2002-1 to 2002-16 in the ½ gain stageare all controlled by the same AGC signal 1924 _(—)½. Likewise, the unitgain amplifiers 2002-17 to 2002-24 in the ¼ gain stage are allcontrolled by the same AGC signal 1924 _(—)¼, and so on. Therefore, atthe 0-gain threshold, all of the amplifiers 2002 in a binary weightedgain stage stop contributing to the gain of the LNA 304, almostsimultaneously.

The variable current source 2122 provides current bias for thedifferential pair 2116. The variable current source 2122 is controlledby the bias control signal 1926 from the control block 1930 (FIG. 3). Inone embodiment of the invention, the variable current source 2122 isshut-off when the 0-gain threshold is reached, and all the differentialpair output current 2117 has been steered to the analog power supplyav_(dd). This reduces overall signal distortion in the output signal 305because non-linear feedthrough is removed from the signal path.Preferably this is done only in the binary weighted amplifiers of thehigh gain amplifier 1918, but could also be done it the low gainamplifier 1922, if so desired.

Referring back to FIG. 19, the overall AGC control for the LNA 304 isnow discussed. When the input differential RF signal 303 has arelatively large amplitude, the master AGC control signal 352 reducesthe gain of the LNA 304. More specifically, the DC amplifiers 1923receive the AGC control signal 352 and generate individual AGC controlsignals 1924 to reduce the gain of the binary weighted gain stages inthe high gain and low gain amplifiers 1918 and 1920. The gain of thehigh gain amplifier gain stages are reduced first and then the gain ofthe low gain amplifier gain stages are reduced. More specifically, the ½gain stage is reduced first until it reaches the 0-gain threshold, andthen the ¼ gain stage is reduced, and so on. Eventually, all the binaryweighed gain stages in the high gain amplifier 1918 stop contributinggain to the LNA 304. If further gain reduction is required, then thebinary weighted gain stages are reduced in the low gain amplifier 1922.The practice of fading the gain on the high amplifier 1918, and thenfading the gain of the low gain amplifier 1922, produces less than 1 dBof noise figure reduction for each 1 dB of attenuation as the gain isreduced. The 12 dB attenuation value for the attenuator 1902 isdetermined to provide a good comprise between noise figure anddistortion. In embodiments of the invention, the gain ranges of the highgain amplifier 1918 and the low gain amplifier 1922 overlap. In otherwords, the gain of the low gain amplifier 1922 drops before all the gainis gone in the high gain amplifier 1918.

In one embodiment, the control block 1930 also powers down theamplifiers 2002 in the high gain amplifier 1918 using the bias controlsignals 1926 when all of the binary weighed gain stages in the high gainamplifier 1918 are providing 0 gain. (i.e. all the output current 2117is being steered to av_(dd) in each of the ½ gain stage, ¼ gain stage,and ⅛ gain stage) It has been found that this improves signal distortionin the LNA output signal 305 because it removes non-linear signalfeedthrough parasitic components associated with a powered-up unit gainamplifier 2002. Preferably, the high gain amplifier 1918 is powered downonly when all the binary weighed stages in the amplifier 1918 are at the0 gain threshold. Alternatively, the binary weighed stages in the highgain amplifier 1918 could be turned-off one at a time, as they reach the0-gain threshold. For example, the ½ gain stage could be powered downwhen it reaches the 0-gain threshold, without waiting for the ¼ gainstage to reach the 0-gain threshold. Preferably, the binary weightedgain stages in the low gain amplifier 1922 remain powered-up, even afterthey have reached the 0-gain threshold. This can be done because the lowgain amplifier 1922 produces little signal distortion. (i.e. thenon-linear parasitics are relative small because of the smaller gain)However, these binary weighted gain stages in the low gain amplifier1922 could also be powered-down at 0-gain, if so desired.

FIG. 22 further illustrates the output matching network 1928 thatmatches the output of the LNA 304 to the input of the differentialmixers 334. The output matching network 1928 includes series inductors2202 a and 2202 b, inductors 2204 a and 2204 b, resistors 2208 a and2208 b, capacitors 2206 a and 2206 b, feedforward capacitors 2210 a and2210 b, and feedforward resistors 2212 a and 2212 b. In embodiments, theinductors 2202 are 8.8 nH inductors, the inductors 2204 are 6.5 nHinductors, the capacitors 2206 are 4.55 pF “finger” capacitors, theresistors 2208 are 44.4 ohm resistors to av_(dd), the feedforwardcapacitors 2210 a are 0.1 pF and the feedforward resistors 2212 are 1.5Kohms. These component values provide a good impedance match over thefrequency band of interest (950-2150 MHz).

The DC amplifiers 1923 that generate the AGC control signals 1924 arepreferably DC differential amplifiers. These differential amplifiers canhave a resistor across their inputs to reduce (or degenerate) the signalgain. Furthermore, a DC amplifier 1929 generates the bias controlsignals 1926 that power-up and power-down the binary weighed gain stagesin the high gain amplifier 1918 and the low gain amplifier 1922. The DCamplifier 1929 preferably does not have the degeneration resistor acrossits inputs. Therefore, the high amplifier is quickly powered-down whenthe 0 gain condition is met for all the gain stages in the highamplifier 1918.

In embodiments, the DC amplifier 1923 that controls the unit gainamplifier 2002-35 in the 1/32 gain stage amplifier can be configured foreither variable gain control or always high gain, using the I²C bus 354.

CMOS Twisted Pair Lines

In embodiments of the invention, the transmission lines that connect oneor more of the circuit elements in the tuner 306 are twisted pair linesthat are implemented on the CMOS substrate. For example, in the LOgeneration circuit 308, the differential lines 313 a and 313 b can beCMOS twisted pair lines. Furthermore, the CMOS twisted pair lines can beused in other parts of the tuner 306.

FIGS. 23 and 24 further illustrate CMOS twisted pair lines 2300. FIG. 23illustrates a top view of the twisted pair 2300, which includes atransmission line 2302 and a transmission line 2304 that cross eachother at junctions 2306 and 2308. FIG. 24 illustrates an isometric viewof the twisted pair 2300. As shown in FIG. 24, the transmission line2302 includes a first metal 2404 and a second metal 2402. The firstmetal 2404 is deposited on the CMOS substrate, and the second metal 2402is deposited on top of the first metal 2404 to form the transmissionline 2302. Likewise, the second transmission line 2304 includes a firstmetal 2408 and a second metal 2406. The first metal 2408 is deposited onthe CMOS substrate, and the second metal 2406 is deposited on top of thefirst metal 2408 to form the transmission line 2304. At the junction2306, the second metal 2402 of the transmission line 2302 crosses overthe first metal 2408 of the transmission line 2304. Likewise, at thesecond junction 2308, the second metal 2406 of the transmission line2304 crosses over the first metal 2404 of the transmission line 2302.

In embodiments, each section of twisted pair lines has 2 twists, whichequalizes the R and C on each line. For example, in FIG. 23, the twotwists are the crossover at the junctions.

In embodiments, vias (e.g. solid vias) can be added from the first metallayer to the second metal layer to lower the resistance.

The twisted pair lines 2300 improve the differential nature of thedifferential transmission lines when compared to conventionaltransmission lines. More specifically, the characteristic impedance ismore carefully controlled, and impedance-perturbing effects of anynearby metal patterns are reduced. Furthermore, the fields decrease morerapidly with distance away from the twisted pair.

Furthermore, the twisted pair provides self-shielding in conjunctionwith differential inputs and outputs by common mode rejection. Couplingis reduced from nearby lines that run along side for distances greaterthan the twisted length. Furthermore, the current path twists in boththe horizontal and vertical dimensions. Furthermore, the shieldingeffect for differential mode is comparable to actual metal shielding.

The twisted pair shown in FIGS. 23 and 24 can be implemented in CMOS.The symmetrical structure allows for easy layout as a mosaic. Twoseparate twisted pairs can be stacked vertically in standard 5 metalprocesses. Staggered twists or different twist pitch keeps linesisolated. Metal is used to its maximum effect. In other words, all ofthe cross section is used to pass signal current with low resistance.

Furthermore, the CMOS twisted pair reduces or eliminates the need forshielding metal on the sides or bottom of the circuit. These shields addunwanted capacitance and lowers the characteristic impedance of thesubstrate.

Conclusion

Example embodiments of the methods, systems, and components of thepresent invention have been described herein. As noted elsewhere, theseexample embodiments have been described for illustrative purposes only,and are not limiting. Other embodiments are possible and are covered bythe invention. Such other embodiments will be apparent to personsskilled in the relevant art(s) based on the teachings contained herein.Thus, the breadth and scope of the present invention should not belimited by any of the above-described exemplary embodiments, but shouldbe defined only in accordance with the following claims and theirequivalents.

1. A local oscillator generation circuit, comprising: a plurality ofvoltage controlled oscillators configured to generate a plurality ofdifferential local oscillator (LO) signals that each cover a differentfrequency range; a plurality of amplifiers coupled to respective outputsof said VCOs, wherein a VCO is selected by enabling one or more of saidamplifiers that correspond to said selected VCO to provide an output LOsignal, and disabling more one or more amplifiers that do not correspondto said selected VCO; a LO correction circuit that is configured toadjust an amplitude level of said output LO signal based on a controlsignal, said control signal based on measuring said amplitude level ofsaid output LO signal.
 2. The local oscillator generation circuit ofclaim 1, wherein said means for generating further comprises a pluralityof polyphase circuits that correspond to said plurality of localoscillator signals, each polyphase circuit configured to generate saidoutput LO signal with in phase (I) and quadrature (Q) components.
 3. Thelocal oscillator generation circuit of claim 1, wherein said LOcorrection circuit comprises: a variable amplifier that variablyamplifies said output LO signal according to said control signal; and alevel detect circuit, connected to the output of said variableamplifier, that generates said control signal based on an output levelof said variable amplifier.
 4. The local oscillator generation circuitof claim 3, wherein said variable amplifier includes a field effecttransistor (FET) configured so that said control signal controls acurrent of said FET, and thereby a gain of said variable amplifier. 5.The local oscillator generation circuit of claim 1, further comprising afeedback loop coupled across said plurality of VCOs.
 6. The localoscillator generation circuit of claim 5, wherein said feedback loopcompares a feedback signal with a reference signal.
 7. The localoscillator generation circuit of claim 6, wherein said feedback loopincludes a frequency divider that provides the feedback signal based onsaid output LO signal.
 8. The local oscillator generation circuit ofclaim 6, wherein said feedback loop further includes a phase detectorthat compares said feedback signal with said reference signal.
 9. Thelocal oscillator generation circuit of claim 8, wherein said feedbackloop further includes a charge pump driven by an output of said phasedetector.
 10. The local oscillator generation circuit of claim 9,wherein said feedback loop further includes a loop filter coupled to anoutput of said charge pump, and wherein an output of said loop filtertunes a frequency of said selected VCO.
 11. A local oscillatorgeneration circuit, comprising: means for generating a plurality oflocal oscillator (LO) signals that each cover a different frequencyrange; means for selecting an output LO signal from said plurality ofdifferential LO signals; and a LO correction circuit that is configuredto adjust an amplitude level of said output LO signal based on a controlsignal, said control signal based on measuring said amplitude level ofsaid output LO signal, said LO correction circuit including, a variableamplifier that variably amplifies said output LO signal according tosaid control signal; and a level detect circuit, connected to the outputof said variable amplifier, that generates said control signal based onan output level of said variable amplifier.
 12. A local oscillatorgeneration circuit, comprising: a plurality of voltage controlledoscillators configured to generate a plurality of differential localoscillator (LO) signals that each cover a different frequency range; aplurality of amplifiers coupled to respective outputs of said VCOs,wherein a VCO is selected by enabling one or more of said amplifiersthat correspond to said selected VCO to provide an output LO signal; anda LO correction circuit that is configured to adjust an amplitude levelof said output LO signal based on a control signal, said control signalbased on measuring said amplitude level of said output LO signal; saidLO correction circuit including, a variable amplifier that variablyamplifies said output LO signal according to said control signal; and alevel detect circuit, connected to the output of said variableamplifier, that generates said control signal based on an output levelof said variable amplifier.